Resonant power converter for radio frequency transmission and method

ABSTRACT

A resonant power converter for ultra-efficient radio frequency transmission and associated methods. In one exemplary embodiment, the invention is digitally actuated and uses a combination of a noise-shaped encoder, a charging switch, and a high-Q resonator coupled to an output load, typically an antenna or transmission line. Energy is built up in the electric and magnetic fields of the resonator, which, in turn, delivers power to the load with very little wasted energy in the process. No active power amplifier is required. The apparatus can be used in literally any RF signal application (wireless or otherwise), including for example cellular handsets, local- or wide-area network transmitters, or even radio base-stations.

PRIORITY

This application is a continuation of prior U.S. patent application Ser.No. 10/382,297 filed on Mar. 4, 2003 which issued as U.S. Pat. No.7,924,937 on Apr. 12, 2011 which claims benefit of U.S. ProvisionalPatent Application Nos. 60/361,812 filed Mar. 4, 2002 and 60/361,813filed Mar. 4, 2002.

RELATED APPLICATIONS

This application is related to co-owned and U.S. patent application Ser.No. 10/382,326, now U.S. Pat. No. 7,525,455, entitled “CODER APPARATUSFOR RESONANT POWER CONVERSION AND METHOD” filed Mar. 4, 2003, whichclaims priority benefit of U.S. provisional patent application Ser. No.60/361,813 of the same title filed Mar. 4, 2002, both of which areincorporated by reference herein in their entirety.

FIELD OF THE INVENTION

The present invention relates generally to radio frequency signals, andspecifically to apparatus and methods for radio frequency (RF) signaltransmission, reception, and/or modulation.

DESCRIPTION OF RELATED TECHNOLOGY

As is well understood, the so-called “PAE” (Power Added Efficiency) ofan amplifier is simply the power output delivered to a load, divided byto the DC input power required for producing such amplification.Typically, for the exemplary CDMA cellular handset, the PAE is about 33%at the maximum transmit power level, and less than 10% on the average.Hence, if a power amplifier having a PAE of 33% puts out 1 Watt of RFpower, it requires 3 Watts of DC battery power, effectively dissipating2 Watts of heat in the process. This is obviously less than optimalusage of batter battery power.

Typical prior art RF transmission systems utilize methods consisting oneor more of the following steps in series: (1) modulating digital datastreams independently into an in-phase and quadrature vector pair, oralternatively into an amplitude and phase vector pair; combining thevectors at some later subsequent point in the signal conversion process;(2) digitally filtering the data vectors; (3) converting the filteredvectors into analog form through D/A conversion; (4) up-converting theD/A outputs into RF signal vectors through one or more stages ofmodulation by an RF oscillator followed by image reject filtering foreach up-conversion stage; (5) pre-amplifying (or variable gainamplifying) the output of the final up-conversion stage; (6) amplifyingthe preamplifier output signal with a power amplifier, wherein theamplifier is a typically a Class-A or Class-AB type.

Alternatively, U.S. Pat. No. 6,181,199, to Dent, et al. entitled “PowerIQ modulation systems and methods” discloses an RF transmission methodwhich differs from the aforementioned methods in that the amplitude ofthe signal vectors are used to amplitude modulate the power supply of aClass-C or Class-D switching power amplifier, which also receive phaseinformation of the signal vector as one of its inputs. U.S. Pat. No.6,198,347 to Sander, et al. entitled “Driving circuits for switch modeRF power amplifiers,” describes a means of driving such an amplifier. Ineither of the aforementioned cases, a signal which has already beenconverted into the analog RF domain is amplified into a load.

U.S. Pat. No. 5,353,309 to Agazzi and Norsworthy entitled “ISDNtransmitter,” discloses a means of using a digital delta-sigma modulatorfor baseband digital telephone transmission. The delta-sigma modulatoroutput is connected to an active analog lowpass filter, and the lowpassfilter drives an active power amplifier, which in turn drives an ISDNtelephone transmission line. There are no means taught in the aforesaiddisclosure, however, for generating enough power to drive the linedirectly without the need for an active power amplifier. Also, theaforesaid disclosure does not provide any means for converting power toan RF carrier frequency, as the described system is a low frequencybaseband transmission system rather than an RF passband transmissionsystem.

U.S. Pat. No. 5,701,106 to Pikkarainen et al. entitled “Method andmodulator for modulating digital signal to higher frequency analogsignal” discloses a baseband digital signal in (I, Q) taken to adelta-sigma digital-to-analog converter sampled at an intermediatefrequency and converted to analog for further up-conversion in theanalog domain to an RF carrier frequency. No means for directlyup-converting the digital baseband signal to the RF carrier frequency,nor for converting DC power directly to RF carrier power, are disclosed.

U.S. Pat. No. 6,321,075 to Butterfield entitled-“Hardware-efficienttransceiver with delta-sigma digital-to-analog converter” is similar tothe invention of the aforementioned U.S. Pat. No. 5,701,106 inasmuch asa delta-sigma modulator in (I,Q) is taken to form an intermediatefrequency and then converted to analog for further up-conversion in theanalog domain to an RF carrier frequency.

FIGS. 1 a-1 c are illustrative of the various prior art architecturesdescribed above.

IEEE Press article number 0-7803-6540 by Keyzer et al, “Digitalgeneration of RF Signals for Wireless Communications With Band-PassDelta-Sigma Modulation” and incorporated by reference herein (“Keyzer”),describes a wireless transmitter incorporating bandpass delta-sigmamodulation used in conjunction with a switching mode power amplifier.See also “Generation of RF Pulsewidth Modulated Microwave Signals UsingDelta-Sigma Modulation” by Keyzer, et al., IEEE Publication 0-7802-7239dated May 2002.

In full-duplex frequency-division systems such as CDMA, the transmitterand receiver can both be on simultaneously. When the transmitter is on,it typically generates noise or distortion that can fall into thereceive band. Using the CDMA standards of IS-95, IS-95a, IS-98, orIS-2000 as examples, the receive band is 80 MHz offset from the transmitband. A typical power amplifier used in a CDMA handset transmittergenerates a noise density of approximately −135 dBm/Hz in the receiveband. For example, for the North American PCS band of operation, thereceive band is 80 MHz higher than the transmit carrier frequency. Inorder to prevent degrading the receiver's sensitivity, the noisegenerated from the transmitter needs to be suppressed to a level belowthe thermal noise floor of the receiver. The thermal noise floor of thereceiver is approximately −174 dBm/Hz. Therefore, the difference betweenthe noise floor of the power amplifier and the thermal noise floor isgreater than 40 dB. This level of noise suppression is commonly achievedthrough a duplexer, the construction and operation of which is wellknown to one of ordinary skill in the art. A duplexer is a 3-portdevice, with one port connected to the output of the power amplifier,another port connected to the antenna, and a third port connected to thereceiver's input.

A particularly difficult problem in using noise-shaping encoders infull-duplex transceivers is that high levels of quantization noise aregenerated out of band, and this may even further corrupt the receiveband. The Keyzer reference described above alludes to this problem, yetdoes not (i) recognize just how severe the problem would be using theirdisclosed approach, or (ii) suggest a solution to meeting therequirements in a practical system. Keyzer uses a second-order bandpassdelta-sigma modulator operating at 4 times the carrier frequency, F_(c),yet doesn't consider the noise floor in the adjacent receive band underthese conditions. The assignee hereof performed a simulation for thepurposes of measuring the quantization noise generated by the bandpassdelta-sigma modulator of Keyzer. Conservative assumptions were made withrespect to the maximum power level required in the relevant standards,and the available battery technology available in typical handsets.Based on these assumptions, a simulation model was constructed anddetermined that the quantization noise would be on the order of −94dBm/Hz at the antenna, without any analog filtering or duplexersuppression. This means that in order to suppress this level ofquantization noise below the thermal level of −174 dBm/Hz, more than 80dB of filtering is required. Furthermore, 80 dB of suppression is needednot just at a single frequency, but over the entire receiver band. Forthe exemplary North American PCS band of CDMA, this bandwidth is 60 MHz,covering 1.93-1.99 GHz. If an analog RF filter with 80 dB of suppressionover this whole frequency range existed in the current state-of-the-art,it would necessarily have a high insertion loss, and would also becomparatively expensive: Therefore, both efficiency and economy aresignificantly impacted using this approach. Thus, there exists a needfor a more efficient and economical solution to suppressing the noise inthe receiver band.

Another particularly difficult problem not addressed by the invention ofKeyzer relates to the extremely high sample rate of the bandpassdelta-sigma modulators. At PCS frequencies, the Keyzer scheme wouldrequire arithmetic logic circuits and registers inside the delta-sigmamodulator operating at nearly 8 GHz. In a portable battery-powered PCShandset, the power consumption just for the bandpass modulator logicwould be enormous in any practical semiconductor technology available atthe time of this writing, an in fact any currently envisioned. Thus,there also exists a need to reduce the clock rate of the delta-sigmamodulators.

A third difficult problem not addressed by Keyzer relates to theinterface between the switch-mode power amplifier and the analog filterthat follows. Specifically, no disclosure or teaching on how to build anoperative switch mode amplifier, and efficiently “drive” it into afilter, is provided. Furthermore, no specific coupling architecture issuggested or described. Hence, a significant technological challenge isleft un-addressed by Keyzer.

It is further noted that Keyzer teaches or infers nothing regarding anyinterpolation filters, or any specific implementation(s) ofinterpolation filters.

Relevant discussion on the subject of delta-sigma data conversion may befound in the textbook by Norsworthy, et al. entitled Delta-Sigma DataConverters, IEEE Press, 1997. In Chapter 9 (beginning on p. 282), thesubject of bandpass delta-sigma modulators is addressed. Bandpassdelta-sigma A/D converters have been used for intermediate frequency(IF) demodulation since the early 1990's. However, no disclosurerelating to the use of delta-sigma D/A converters for radio transmissionor RF power conversion is provided.

As evidenced by the foregoing, application of the fundamental conceptsof delta-sigma modulation has not produced tenable solutions to theaforementioned problems. Thus, there remains a salient need for animproved apparatus and method for converting signals digitally into RFpower without having to build an active amplifier, and without having tofirst convert a digital data signal into the analog domain at afrequency substantially less than the RF carrier frequency. Suchimproved apparatus and methods would also inherently provide a highdegree of power efficiency to reduce power consumption and, inter alia,thereby increase the battery longevity of wireless handsets.

SUMMARY OF THE INVENTION

The present invention satisfies the aforementioned needs by providing animproved method and apparatus for signal and power conversion.

In a first aspect of the invention, an improved apparatus for resonantpower conversion of RF signals is disclosed. The apparatus generallycomprises a pulse input source adapted to generate a plurality ofpulses; a resonator operatively coupled to the pulse input source; and atransmission medium operatively coupled to the resonator's output andadapted to transmit a plurality of RF signals. In one exemplaryembodiment, the resonator has a resonant frequency at or substantiallynear a carrier frequency, and is further adapted to efficiently storeenergy (through, inter alia, selective reinforcement of at leastportions of a plurality of the generated pulses) for subsequenttransmission thereof. Specifically, one variant uses a digitallyactuated resonant power (DARP) converter comprising a noise-shapingencoder for receiving digital data at a clock rate F_(c)/L₁, where L₁ isa multiple or sub-multiple of a carrier frequency F_(c), and encodingthe digital data. A power supply having a frequency at or substantiallynear DC is also provides, as well as a load impedance coupled to theresonator for receiving energy stored in the resonator. A chargingswitch is coupled to the noise-shaping encoder, power supply, resonator,and a clock having a clock rate L₂F_(c), where L₂ is a multiple of thecarrier frequency F_(c). The charging switch is adapted to: (i) receiveencoded data from the noise-shaping encoder; (ii) sample the voltage orcurrent of the power supply; and (iii) deliver the power supply voltageor current samples to the resonator.

In a second aspect of the invention, an improved method for performingresonant power conversion is disclosed. The method generally comprisesgenerating a plurality of pulses; inputting the pulses into a resonatoradapted to selectively reinforce at least portions of a plurality of thepulses; selectively reinforcing the aforementioned portions of thepulses; and transmitting the selectively reinforced signals over atransmission medium.

In a third aspect of the invention, an improved transfer function fornoise-shaping encoder implementation is disclosed which allows thefunction to be implemented with two lowpass encoders operating at alower sampling rate.

In a fourth aspect of the invention, an improved noise-shaping encoderapparatus is disclosed. In one exemplary embodiment, the improvedencoder apparatus comprises a table lookup and data-addressable memory.

In a fifth aspect of the invention, improved methods and apparatus forcontrolling power gain of the apparatus are disclosed. In one exemplaryembodiment, power gain is controlled completely in the digital domain.In a second embodiment, gain is controlled by a combination of digitaland analog means.

In a sixth aspect of the invention, an improved charging switchapparatus is disclosed. In one exemplary embodiment, charging switchesare implemented on the same semiconductor substrate as the noise-shapingencoder logic in a manner that significantly reduces power consumption.

In a seventh aspect of the invention, an improved resonator andtransformer combination is disclosed, including improved dynamicimpedance termination.

In an eighth aspect of the invention, an improved resonator apparatus isdisclosed, wherein the resonator is combined as part of a duplexer.

BRIEF DESCRIPTION OF THE DRAWINGS

The features, objectives, and advantages of the invention will becomeapparent from the detailed description set forth below when taken inconjunction with the drawings, wherein:

FIGS. 1 a, 1 b, and 1 c are functional block diagrams illustrating thegeneral configuration of typical prior art RF transmitter and poweramplification systems.

FIG. 2 is a functional block diagram illustrating the generalconfiguration of an exemplary converter apparatus according to thepresent invention.

FIG. 2 a is a functional block diagram illustrating an exemplaryconfiguration of the converter apparatus of FIG. 2.

FIG. 3 is a functional block diagram illustrating one exemplaryembodiment of the converter apparatus of FIG. 2.

FIG. 4 is a functional block diagram illustrating another exemplaryembodiment of the converter apparatus of FIG. 2.

FIG. 5 is a graphical representation of exemplary noise spectra (a)before, and (b) after, input to the noise-shaping encoder(s) of theapparatus of FIG. 3.

FIG. 6 is a graphical representation of a noise transfer function (fullduplex transceiver) associated with an exemplary third-ordernoise-shaping encoder according to the invention.

FIG. 7 is a graphical representation of first exemplary noise spectrafor L₁=1 and L₂=4 (a) before the digital quadrature modulator, (b) afterthe digital quadrature modulator but before the resonator, and (c) afterthe resonator, of the apparatus of FIG. 3.

FIG. 8 is a graphical representation of second exemplary noise spectrafor L₁=2 and L₂=4 (a) before the digital quadrature modulator, (b) afterthe digital quadrature modulator but before the resonator, and (c) afterthe resonator, of the apparatus of FIG. 3.

FIG. 9 is a graphical representation of third exemplary noise spectrafor L₁=1 and L₂=8 (a) before the digital quadrature modulator, (b) afterthe digital quadrature modulator but before the resonator, and (c) afterthe resonator, of the apparatus of FIG. 3.

FIG. 10 is a functional block diagram of an alternate embodiment of theapparatus of FIG. 2, illustrating an exemplary configuration of atermination network on the switch side of the resonator.

FIG. 11 is a functional block diagram of an alternate embodiment of theapparatus of FIG. 2, illustrating an exemplary scheme for incorporatingthe resonator into a duplexer.

FIG. 12 is a graphical representation of the time-domain response of anexemplary resonator configuration of the invention to a short sequenceof switch excitations.

FIG. 13 is a graphical representation of exemplary spectral output fromthe final stage of interpolation filtering at 1536 times the symbol rate(images suppressed as previously discussed)

FIG. 14 is a graphical representation of exemplary spectral output fromthe encoder, at the input to the switches.

FIG. 15 is a graphical representation of exemplary frequency response ofthe resonator.

FIG. 16 is a graphical representation of exemplary spectral output fromthe resonator over the full measurable band.

FIG. 17 is a graphical representation of exemplary spectral output fromthe resonator over the PCS transmit band.

FIG. 18 is a graphical representation of exemplary transmit leakagepower into the PCS receive band with an 80 MHz offset.

FIG. 19 is a graphical representation of exemplary time domain outputfrom the resonator.

FIG. 20 is a graphical representation of exemplary phase response of anideal case (i) without delta-sigma modulation, and (ii) with delta-sigmamodulation.

FIG. 21 is a graphical representation illustrating the difference intime domain response from simulations (generated via Cadence DesignSystems, Inc. SPICE program) of the exemplary switch/resonator circuitof the invention using (i) actual production-quality GaAs MESFET models,and (ii) an ideal switch.

DETAILED DESCRIPTION OF THE INVENTION

Reference is now made to the drawings wherein like numerals refer tolike parts throughout.

As used herein, the terms “transmit”, “transmission” and “transmitting”for convenience may generally be considered to refer to both the acts oftransmitting signals and receiving signals, as applicable.

As used herein the terms “memory” and “storage device” are meant toinclude any means for storing data or information, including, withoutlimitation, RAM (e.g., SRAM, SDRAM, DRAM, SDRAM, EDR-DRAM, DDR), ROM(e.g., PROM, EPROM, EEPROM, UV-EPROM), magnetic bubble memory, opticalmemory, embedded flash memory, etc.

It will be recognized that while the following discussion is castprimarily in terms of a wireless RF handset (e.g., cellular telephones),the present invention is in no way limited to any particular wirelessmethod, air interface, or architecture, or for that matter wirelessapplications. The invention may be applied with equal success tonon-wireless systems of any kind as well, consistent with anylimitations described herein.

Referring now to FIGS. 2 and 2 a, an exemplary generalized form of theresonant power converter of the present invention is described. As shownin FIG. 2, the apparatus 200 of the invention can be generally thoughtof as a pulse input source 202 whose output is coupled to the input of aresonator 204, the output of the resonator coupled to a load ortransmission means (e.g., wireless antenna, transmission line, etc.)206. The particular properties of the resonator of the invention aredescribed in greater detail subsequently herein. One highly advantageousattribute of the configuration of FIG. 2 (and in fact, all otherdisclosed embodiments of the invention) is that no amplifier whatsoeveris needed; rather, the resonator 204 acts effectively as an ultra-highefficiency power supply. This not only obviates the cost, complexity,etc. associated with prior art designs having a power amplifier, butalso provides significantly increased efficiency, thereby having greatimpact on, inter alia, the power consumption of RF devices.

FIG. 2 a illustrates an exemplary digitally actuated resonant power(DARP) device 220 based on the generalized model of FIG. 2. This device220 includes a noise shaping encoder 222 and charging switch 224 as partof the pulse data source 202 of FIG. 2.

Referring now to FIG. 3, one exemplary architecture 300 (based on thatof FIG. 2 discussed above) assumes that digital data is to be passbandmodulated, and then transmitted at radio frequencies to a loadimpedance, such as an antenna or transmission line. Digital data iscoupled to one or more noise-shaping encoders 302 that spectrally shapequantization noise, pushing the noise out of the band of interest. Theoutput of the noise-shaping encoder 302 is typically one or a few bitswide, and the output word rate of the encoder is typically a multiple orsub-multiple L₁ of the RF carrier frequency F_(c). The output of theencoder 302 is coupled to at least one charging switch 304. Suchcharging switch(es) may comprise any number of different configurationswell known to those of ordinary skill in the electronic arts, asdiscussed in greater detail below. The purpose of the charging switch304 is to sample a DC power supply voltage VDC (or alternate lowfrequency power supply) and rapidly (e.g., instantaneously) switchcharge onto the internal capacitance of the resonator 306, at either thepositive phase of the RF carrier frequency F_(c), or at the anti-phaseof F_(c) 180 electrical degrees later. A moment after the resonator'scapacitance is charged, current begins to flow through the inductance ofthe resonator, and the resonator begins to oscillate at F_(c). Theresonator 306 is in the illustrated embodiment assumed to be a high-Qresonator to effectively multiply the load impedance Z_(L) 309 at theoutput of the resonator 306, as seen by the input to the resonator. Theload impedance 309 may either be an antenna, transmission line, or othersimilar modality, although other forms of impedance may also besubstituted. The charging switch(es) 304 continues to actuate theresonator at either the positive phase or antiphase of F_(c). The outputsamples {i_(k), q_(k)} of the noise-shaping encoder(s) 302 determine thevalue of the charging switch samples at each sampling instant.

The noise-shaping encoder 302 may either be implemented as two lowpassencoders (FIG. 3), as a single bandpass encoder 406 (FIG. 4), or inother equivalent configurations. In applications where power consumptionis a major concern and where the clock frequencies are near the upperlimit of the “available” semiconductor technology (here, “available”meaning available limited by the state of the art, or alternativelylimited by other constraints such as target cost, IC operating voltage,die size, etc.), there may be a significant advantage in implementingthe noise-shaping encoder as two lowpass encoders at a lower clockfrequency, as opposed to a bandpass encoder at a higher clock frequency.

The noise-shaping encoder function of the present invention mayalternatively be implemented as a table lookup function, where thenoise-shaping encoding is performed for example off-line in advance ofthe known finite states of the digital data. For example, the resultsmay be stored in a data addressable memory or storage device forconvenience. Myriad other options exist. Such configurations and optionswill be readily recognized and understood by those of ordinary skill inthe signal processing arts, and accordingly are not described furtherherein.

The embodiments of FIGS. 3 and 4 are now described in greater detail.Beginning with a digital processor 310 (which may comprise a DSP, RISCprocessor, CISC processor, ASIC, or the like having sufficientcapability), a digital data stream is separated into two orthogonalsignal vectors: an in-phase vector (I) and a quadrature vector (Q), at asymbol rate of F_(b). The digital data can be one of any type of knownmodulation formats including, without limitation, shift keying (e.g.,π/4 QPSK, FSK, GFSK, GMSK, etc.), amplitude modulation (e.g., QAM,etc.), and the like. The binary data is treated as a 1 for the “one”state and a −1 for the “zero” state, although it will be recognized thatthese designations are arbitrary. Digital interpolation filters 312provide channel filtering on the symbols, and can be one of any type ofknown symbol filters, such as the family of raised cosine filters thatprovide zero intersymbol interference. For many specific datatransmission standards, such as IS-2000, the symbol filtering isspecified. In any case, the symbol filtering can be incorporated withinthe overall filtering performed by digital interpolation filters 312.The filters 312 also increase the sampling rate by the ratio(F_(c)/L₁)/F_(b), where F_(c) is the carrier frequency of the desired RFsignal to be transmitted. In the exemplary embodiment, L₁=1, whichsimply means that the lowpass noise-shaping encoders 302 are operatingat a sampling rate equal to the carrier frequency, F_(c). In order tolower the power consumption of the encoder logic 302, it may bedesirable to lower the sampling rate of the encoders (12), therefore,L₁=2, or even larger, could be chosen, at the sacrifice of greaterquantization noise in the desired bands of interest. In practice, theoverall interpolation ratio (F_(c)/L₁)/F_(b) will typically still be alarge number. By way of example for the IS-2000 standard, if the symbolrate F_(b) is 1.23 MHz and if the carrier frequency F_(c) is 1.88 GHz,and if L₁=1, then the overall interpolation ratio is approximately equalto 1529.945. Obviously, other frequencies may be selected, resulting inother interpolation ratios. In general, the interpolation ratio betweenthe carrier rate and the symbol rate will not necessary be an integer,and the baseband symbol clock may come from a completely independent(incommensurate) source with respect to the carrier clock. It will beappreciated that various compatible techniques exist at the time of thiswriting for carrying out fractional sample rate conversion, includingbut not limited to, fractional-N synthesis phaselock loops. Anall-digital technique employing fractional decimation and interpolationmay also be employed, appropriate with the scope and architecture ofthis invention. The reader is referred to, for example, “Sample RateConversion for Software Radio” by Hentschel et al., IEEE CommunicationsMagazine, August 2000, p. 142-150, incorporated herein by reference.Many other well known techniques for synchronizing asynchronous clocksmay be applied to this invention by one with ordinary skill in the artgiven the present disclosure.

The digital interpolation filters 312 of FIG. 3 may actually be used todistribute the interpolation ratio (F_(c)/L₁)/F_(b) across two or morestages of interpolation for the purpose of reducing the cost and powerconsumption of the implementation by relaxing the multiplication ratesand complexity. The design of multi-rate digital filters is well knownin the literature, and such techniques applied to delta-sigma convertersare also well known in the art, and may be found, e.g., in Chapter 13 ofthe aforementioned textbook by Norsworthy, et al., Delta-Sigma DataConverters, IEEE Press, 1997, incorporated herein by reference. Forexample, consider a symbol-rate filter for IS-2000, and let the overallinterpolation filter be divided into three main sections. Let the firstfilter, operating on the incoming symbols, have the followingconstraints: an FIR structure; operating at 8 times the symbol rate;passband cutoff frequency of 0.48 times the symbol rate; passband rippleof less than 1 dB; stopband cutoff frequency of 0.6 times the symbolrate; stopband attenuation of least 60 dB. These constraints will resultin an FIR filter having at least 160 taps, found by using the well-knownRemez exchange algorithm. The second stage of interpolation filteringmay be a sinc-cubic filter having an oversampling ratio of 24, producingimages at multiples of 8 times the symbol rate that are suppressed by atleast 70 dB. A third stage of interpolation filtering may simply be azero-order hold, having an oversampling ratio equal to 8, producingimages at multiples of 8*24=192 times the symbol rate that aresuppressed by at least 50 dB. Thus, if we multiply the oversamplingratios of all three of these filter stages, we have 8*24*8=1536, and theoverall oversampling ratio (F_(c)/L₁)/F_(b) is achieved efficiently. Afractional sampling rate converter may be incorporated within one ormore stages of interpolation, in order to synchronize the basebandsymbol rate to the carrier rate. It will be appreciated that while thisexample embodiment describes one efficient way of achieving theinterpolation filters 312, there are certainly other substitutions thatcan be recognized and made by one of ordinary skill in the art, such asbut not limited to, substituting IIR filters for FIR filters, makingmore stages or fewer stages of interpolation filtering, choosingdifferent relative interpolation ratios, choosing a larger value of L₁in order to lower the interpolation ratio, etc.

The outputs of the interpolators 312 are coupled to the inputs of thenoise-shaping encoders 302. These encoders are separate lowpass encodersin I and Q in the embodiment of FIG. 3, and are effectively up-convertedand combined by digital quadrature modulator 316, resulting in abandpass signal. Alternatively, a bandpass encoder following thequadrature modulator 316 could be substituted for the two lowpassencoders 302 before the quadrature modulator 316, as is shown in theconfiguration of FIG. 4. In either embodiment, the lowpass encoders aredesigned such that the signal energy from interpolators 312 is preservedin the baseband. FIG. 5 illustrates the spectrum (a) prior to encoding,and (b) at the output of encoders 302. The gray shaded portions 510 ofFIG. 5 illustrate the quantization noise that results from the encodingprocess. The encoder creates quantization noise, but effectively pushesthe quantization noise out of band so that most of the noise liesoutside, between F_(b)/2 and (F_(c)/L₁)/2, advantageously resulting in arelatively high signal-to-noise ratio inside the band of interestbetween 0 and F_(b)/2.

As previously described herein, the quantization noise (generated by atypical noise-shaping encoder operating at sampling frequencies in thevicinity of the carrier frequency) may produce very high levels in theadjacent receive band of a full-duplex transceiver. This may result inserious degradation of the receiver if the noise is not adequatelyremoved. One solution described herein entails placing one or more zerosof the encoder noise transfer function at selected frequencies where thequantization noise needs suppression. A simplest form of a second-orderlowpass delta-sigma modulator has a noise transfer function (NTF) givenbelow:H(z)=1−2z ⁻¹ +z ⁻²This results in two zeros at z=1, or 0 Hz. The zeros can be transformedfrom 0 Hz to another frequency by expressing the NTF as follows:H(z)=1−2 cos(2πf _(z) /f _(s))z ⁻¹ +z ⁻²As an example, suppose the desired zeros are ±80 MHz, and the samplingrate of the encoder is 1.88 GHz. The NTF then becomes:H(z)=1−1.928938z ⁻¹ +z ⁻²Simulation performed by the Assignee hereof shows that the quantizationnoise at 80 MHz±F_(b)/2 is suppressed an additional 36 dB compared withthe standard version of the second-order lowpass modulator having bothzeros at 0 Hz. The tradeoff, however, is that some sort of multipliermust be incorporated within the logic to implement the coefficient,instead of a simple shift to implement a multiply by 2. This NTF nolonger has infinite suppression at 0 Hz, and has limited suppression atthe symbol band edges, ±F_(b)/2. None-the-less, it still may be veryuseful in certain applications where such features are acceptable ordesired.

As another alternative, an exemplary third-order encoder is configuredsuch that a single zero is placed at 0 Hz, and the zeros at ±80 MHz aremaintained as before. In order to keep the encoder stable, poles areplaced in the NTF in addition to the zeros. While this may initiallyseem counter-intuitive, it will be appreciated that the noise-shapingfilter is effectively contained within a quantized feedback loop. Acomprehensive treatment on the design of stable high-order delta-sigmamodulators is well known in the art, and can be found for example inChapters 4 and 5 of the textbook by Norsworthy, et al., Delta-Sigma DataConverters, IEEE Press, 1997, previously incorporated herein.

The exemplary third-order noise-shaping encoder, again assuming asampling rate of 1.88 GHz, having zeros at ±80 MHz and at 0 Hz, has aNTF given by the following relationship:

${H(z)} = \frac{\left( {1 - {1.928938z^{- 1}} + z^{- 2}} \right)\left( {1 - z^{- 1}} \right)}{\left( {1 - {1.427625z^{- 1}} + z^{- 2}} \right)\left( {1 - {0.625422z^{- 1}}} \right)}$The NTF for this encoder is illustrated in FIG. 6. Simulation shows thatthe quantization noise at 80 MHz±F_(b)/2 is suppressed an additional 26dB compared with the standard second-order lowpass modulator having bothzeros at 0 Hz. This is about 10 dB less suppression when compared withthe second-order modulator with its zeros moved from 0 Hz to ±80 MHz.However, this third-order modulator has the advantage of excellentsuppression at the symbol band edges, ±F_(b)/2, because there is a zeroat 0 Hz. The tradeoff, however, is that at least one multiplier may needto be incorporated within the logic to implement at least onecoefficient. Simulations have shown that the coefficients controllingthe pole locations are not too sensitive to the stability andperformance of the encoder due to roundoff errors, and simple choicescan be found in implementing the pole locations in the z-plane,resulting in short coefficient word lengths that may be implementedwith, for example, simple shift-add schemes. The extra logic requiredfor the implementation of the carefully selected coefficientmultiplications in many instances is relatively trivial. The reader isreferred to Chapter 10 of the aforementioned textbook by Norsworthy, etal., which teaches techniques on the implementation of digitaldelta-sigma modulators.

The out-of-band gain of this exemplary third-order modulator is 1.57,which means that it is relatively stable for one-bit quantization, inaccordance with Chapter 4 of the aforementioned textbook by Norsworthy,et al., where the suggested maximum out-of-band gain for stability isaround 1.5 or slightly greater. This example is only one of many suchnoise-shaping encoders that can be designed, as various substitutionscan be made, including but not limited to, the sampling rate, thelocations of the poles and zeros, and the order of the encoder.

Similarly, a bandpass delta-sigma modulator using thelowpass-to-bandpass transformation z→−z² can be substituted in lieu oftwo lowpass delta-sigma modulators (one in I and one in Q), and theabove example design can be directly transformed if so desired, and usedas an embodiment of bandpass noise-shaping encoder 406 in FIG. 4. Acomprehensive treatment on the design of bandpass delta-sigma modulatorscan be found in Chapter 9 of the aforementioned textbook by Norsworthy,et al.

The noise-shaping encoders 302 may employ either one-bit or multi-bit(n-bit) quantization. Ideally, the encoders 302 should be free ofspurious tones in the spectrum, and it is often necessary to dither theencoders. A comprehensive treatment on dithered delta-sigma modulatorsmay be found in Chapter 3 of the aforementioned textbook by Norsworthy,et al., wherein a stability criteria test for dithered multibitnoise-shaping encoders is provided (p. 130-131). A desirablecharacteristic is that the encoders have fully dithered quantizers,ensuring that the quantization noise is white. A used herein, the term“fully dithered quantizer” refers to the dither fully spanning onequantization interval. For example, for a ternary encoded quantizer, ifthe output levels are {1, 0, −1}, then the corresponding thresholds are{−0.5, 0.5}, the dither interval is therefore also {−0.5, 0.5}, and thedither generator creates pseudo-random values between these outerlimits. The dither is arithmetically added to the input of the internalquantizer of the encoder. While it is often desirable to fully ditherthe quantizer, this limits the usable dynamic range of the encoder, andmay degrade the stability. For virtually all known noise-shapingencoders, a fully-ditherd quantizer will require multibit quantization.

If the encoders 302 use ternary quantization of {1, 0, −1}, and if thecoder is first order, and if the quantizer is fully dithered over {−0.5,0.5}, then the stable input range of the incoming signal is also {−0.5,0.5}. This surprising and elegant result has significant practicalimplications. Specifically, the maximum pulse density coming out of theencoders in this example is 0.5. If the dither range is limited so thatit does not cover an entire quantization interval, then the inputdynamic range could be increased. However, idle channel tones and spikesmay possibly show up in the quantization noise spectrum.

In the foregoing CDMA-based examples, it was shown how a second- orthird-order encoder could be designed to meet certain systemrequirements. For TDMA systems (including for example GSM), or othersystems with less stringent out-of-band noise requirements, it may bepossible to use first-order encoders. In systems such as TDMA, where thereceiver and transmitter are not on at the same time, the suppression ofquantization noise outside the transmit band of interest is not nearlyas critical. In fact, a more trivial delta-sigma modulator may beemployed. The simplest known delta-sigma modulator is a first-ordermodulator. There are good reasons why one would consider the encoders302 to be the lowest order possible, in order to keep the quantizationnoise from rising too sharply out of band. First-order delta-sigmamodulators have historically been avoided altogether since the inventionof second- and higher-order modulators in the early 1980's. In nearlyall known commercial applications, first-order modulators are avoidedbecause of their inherent generation of high levels of spurious tones,which cause them to be virtually unusable in many practical systemdesigns. A first-order encoder causes the quantization noise to rise atonly 9 dB/octave, while a second-order encoder's noise will rise at 15dB/octave, and a third-order encoder's noise will rise at 21 dB/octave.The passive resonators (FIGS. 10, 11) must attenuate the out-of-bandquantization noise, and the resonators should ideally have a low-ordercharacteristic in order to keep the insertion losses as low as possible.For example, a single-section bandpass resonator will have anattenuation of 6 dB/octave on each side of the resonant frequency.Meeting a 9 dB/octave rise in quantization noise with a 6 dB attenuationfrom the resonator will still cause the quantization noise to rise a net3 dB/octave. If the oversampling ratio is 2048, then there areapproximately 11 octaves of net rise at 3 dB/octave, which may result intoo much out-of-band noise and not meeting the spectral requirements forthe end-system. Hence, another means of attenuating the out-of-bandquantization noise is introduced in the exemplary sample-and-holdinterpolators 318 (FIG. 3), which introduce spectral zeros at DC and atmultiples of 4F_(c), effectively alleviating the problem of inadequateattenuation from the resonator alone, and rolling off the out-of-bandenergy at a dramatic rate.

Referring again to FIG. 3, the outputs of noise-shaping encoders 302 arecoupled to the inputs of sample-and-hold interpolators 318, having aninterpolation ratio of the product of L₁ and L₂. The purpose of thesesample-and-hold interpolators 318 is to interface the output sample rateof the noise-shaping encoders with the modulation frequency of thedigital quadrature modulator 316.

Several examples will be provided below that help describe the operationand behavior of the DARP converter apparatus 300 of FIG. 3.

Example 1

Referring back to the embodiment of FIG. 3, we choose L₁=1 and L₂=4. Wealso set the phase offset, θ=0, inside the cos( ) and sin( ) argumentsdriving the I- and Q-modulators. Data sample {i_(k)} is the kth samplecoming from the in-phase encoder 302 a, and similarly, data sample{q_(k)} is the kth sample coming from the quadrature encoder 302 b.Using a clock at a rate 4F_(c), which is four times higher than thecarrier frequency F_(c), samples from the encoders are effectivelysampled and held by interpolators 318 four successive times before thenext kth sample arrives. I- and Q-modulators 320 are effectivelyarithmetic multipliers. The multipliers each have two inputs and oneoutput. I-modulator 320 a receives the in-phase samples frominterpolator 318 a, and also receives a periodic sequence {1, 0, −1, 0},which is the result of the trigonometric operation cos(2πn/4), thein-phase version of the carrier frequency F_(c) having four samples percarrier cycle. Similarly, Q-modulator 320 b receives the quadraturesamples from interpolator 318 b, and also receives a periodic sequence{0, 1, 0, −1}, which is the result of the trigonometric operationsin(2πn/4), the quadrature version of the carrier frequency F_(c) havingfour samples per carrier cycle. The result of these operations is thecreation of {i_(k), 0, −i_(k), 0} at the output of the I-modulator 320a, and {0, q_(k), 0, −q_(k)} at the output of the Q-modulator 320 b.Hence, every in-phase encoder sample {i_(k)} at F_(c) is transformedinto a four-phase packet {i_(k), 0, −i_(k), 0} at 4F_(c), and everyquadrature encoder sample {q_(k)} at F_(c) is transformed into afour-phase packet {0, q_(k), 0, −q_(k)} at 4F_(c). The combiner 324 thenproduces the data sequence {i_(k), q_(k), −i_(k), −q_(k)} every carriercycle, or period of F_(c). If the noise-shaping encoders (14) areconstrained to binary quantization, then there are four possible suchdata sequences every carrier cycle: {1,1,−1,−1}, {1,−1,−1,1},{−1,−1,1,1}, {−1,1,1,−1}. Each one of these data sequences representsone of four possible signal constellation points. For a comprehensivetreatment on passband data transmission and phase shift keying (PSK)signal constellations in particular, the reader is referred to the text,Data Communications Principles, by Gitlin et al., Plenum Press, 1992,Chapter 5, beginning on p. 325.

Example 2

As in the previous EXAMPLE 1, L₁=1 and L₂=4. This time, however, thephase offset, θ=π/4, is set inside the cos( ) and sin( ) argumentsdriving the I- and Q-modulators. Note that setting the phase offset toπ/4 is important in some instances. In particular, the CDMA standards ofIS-95, IS-95a, IS-98, and IS-2000 call for π/4 offset QPSK as a mode ofmodulation. It is well known to one of ordinary skill in the art of datamodulation, that using π/4 offset QPSK, as opposed to zero offset, hasthe benefit of reducing the peak-to-average, otherwise known as thecrest factor. Going back to the prior description of operation, datasample {i_(k)} is the kth sample coming from the in-phase encoder 302 a,and similarly, data sample {q_(k)} is the kth sample coming from thequadrature encoder 302 b. Using a clock at a rate 4F_(c), which is fourtimes higher than the carrier frequency F_(c), samples from the encodersare effectively sampled and held by interpolators 318 four successivetimes before the next kth sample arrives. I- and Q-modulators 320 a, 320b are, in the illustrated embodiment, effectively arithmeticmultipliers. The multipliers each have two inputs and one output. TheI-modulator 320 a receives the in-phase samples from the interpolator318 a, and also receives a periodic sequence {1,−1,−1,1}, which is theresult of the trigonometric operation cos(2πn/4+π/4), the in-phaseversion of the carrier frequency Fc. Similarly, the Q-modulator 320 breceives the quadrature samples from the interpolator 318 b, and alsoreceives a periodic sequence {1,1,−1,−1}, which is the result of thetrigonometric operation sin(2πn/4+π/4), the quadrature version of thecarrier frequency. F_(c). (Note that we have taken the sign of the cos() and sin( ) arguments here, ignoring the 1/√{square root over (2)}multiplier, a result of the π/4 offset, for the moment.) The result ofthese operations is the creation of {i_(k), −i_(k), −i_(k), i_(k)} atthe output of the I-modulator, and {q_(k), q_(k), −q_(k), −q_(k)} at theoutput of the Q-modulator. Hence, every in-phase encoder sample {i_(k)}at F_(c) is transformed into a four-phase packet {i_(k), −i_(k), −i_(k),i_(k)} at 4F_(c), and every quadrature encoder sample {q_(k)} at F_(c)is transformed into a four-phase packet {q_(k), q_(k), −q_(k), −q_(k)}at 4F_(c). The combiner 324 then produces {(i_(k)+q_(k)),(−i_(k)+q_(k)), (−i_(k)−q_(k)), (i_(k)−q_(k))}.

If the noise-shaping encoders 302 are constrained to binaryquantization, then there are four possible such data sequences everycarrier cycle: {2,0,−2,0}, {0,2,0,−2}, {−2,0,2,0}, {0,−2,0,2}. Thisresults effectively in the insertion of every other sample being a zero,allowing the resonator and switch to rest or settle between sample hits,and reducing the probability of inter-symbol interference at theswitch-resonator interface.

The spectral relationships for EXAMPLES 1 and 2 are graphicallyillustrated in FIG. 7.

Example 3

As in the previous EXAMPLE 2, L₁=1, L₂=4, and θ=π/4. However, ternaryquantization is used within the noise-shaping encoders 302. Therefore,there are nine possible data sequences every carrier cycle, and theconstellation map would consist of a rectangular arrangement of ninesymbol points at the following I-Q coordinates: (1,0), (1,1), (0,1),(−1,1), (−1,0), (−1,−1), (0,−1), (1,−1), and (0,0). The nine possibledata sequences corresponding to these nine symbol points on theconstellation map would therefore be: {1,−1,−1,1}, {2,0,−2,0},{1,1,−1,−1}, {0,2,0,−2}, {−2,0,2,0}, {−1,−1,1,1}, {0,−2,0,2}, {0,0,0,0}.

Example 4

In this case, L₂=2, but every parameter is the same as in EXAMPLE 2,i.e., L₁=4 and θ=π/4. This effectively lowers the sampling rate of theencoders 302 by a factor of two to F_(c)/2 and makes the sample/holdinterpolation ratio, L₁L₂=8, so that the output rate of the sample/holdinterpolators 318 is still 4F_(c) as before. Therefore, one new datasample, {i_(k)} or {q_(k)}, is produced by the noise-shaping encoders302 every eight clock cycles of the digital quadrature modulator 316.The result of these operations is the creation of {i_(k), −i_(k),−i_(k), i_(k)} twice in a row at the output of the I-modulator, and{q_(k), q_(k), −q_(k), −q_(k)} twice in a row at the output of theQ-modulator 320 b. Hence, every I-encoder sample {i_(k)} at F_(c)/2 istransformed into a eight-phase packet {i_(k), −i_(k), −i_(k), i_(k),i_(k), −i_(k), −i_(k), i_(k)} at the output of the I-modulator 320 a ata rate of 4F_(c), and every Q-encoder sample {q_(k)} at F_(c)/2 istransformed into a eight-phase packet {q_(k), q_(k), −q_(k), −q_(k),q_(k), q_(k), −q_(k), −q_(k)} at the output of the Q-modulator 320 b ata rate of 4F. The combiner 324 then produces {(i_(k)+q_(k)),(−i_(k)+q_(k)), (−i_(k)−q_(k)), (i_(k)−q_(k))} twice in a row. If thenoise-shaping encoders 302 are constrained to binary quantization, thenthere are four possible such data sequences every carrier cycle:{2,0,−2,0,2,0,−2,0}, {0,2,0,−2,0,2,0,−2}, {−2,0,2,0,−2,0,2,0},{0,−2,0,2,0,−2,0,2}.

The spectral relationships for this example are graphically illustratedin FIG. 8.

Example 5

It will be recognized that the present invention could be practicedusing any length phase packet or any number of quantization levels inthe noise-shaping encoders. For example, if we desire eight (8) uniquephase states per carrier cycle, then we could set L₁=1 and L₂=8, set thenumber of quantization levels to binary, and set both the digitalquadrature modulator 316 and the charging switches 304 operate at8F_(c). The spectral relationships of this example are graphicallyillustrated in FIG. 9.

The action of the sample/hold interpolators 318 places as sinc(x)function on the spectrum to appear as depicted in FIG. 7( a), placingspectral zeros at multiples of F_(c). The action of the quadraturemodulators 320 effectively shifts the spectrum up to where the passbandis centered at F_(c), as depicted in FIG. 7( b). Two other illustrativeexamples can be seen in FIGS. 8 and 9. In FIG. 8, L₁=2 and L₂=4, whilein FIG. 9, L₁=1 and L₂=8.

While several possible examples have been provided, many other suchcombinations of parameters are also possible within the scope of thisinvention. Therefore, it will be plainly recognized that the inventionis in no way limited only to the foregoing examples. Additional examplesfor other combinations of interpolation ratios, phase packet lengths,sampling rates, constellation maps, and the like, can be readily derivedby one of ordinary skill in the art given the present examples in thisdisclosure.

It will further be recognized that the dither scheme and apparatusdescribed in detail in co-owned and co-pending U.S. patent applicationSer. No. 10/10/382,326, now U.S. Pat. No. 7,525,455, entitled “CODERAPPARATUS FOR RESONANT POWER CONVERSION AND METHOD” filed Mar. 4, 2003,which claims priority benefit of U.S. provisional patent applicationSer. No. 60/361,813 of the same title filed Mar. 4, 2002, previouslyincorporated by reference herein, may be used consistent with thepresent invention. This dither method and apparatus may be applied toliterally any type of encoder of any order (i.e., one through “nth”order), including that described herein, and may utilize any decimationfactor greater than 1 including, e.g., 2, 4, 8, or even non-powers of 2.

The outputs of the quadrature modulators 320 are summed in quadraturewith the combiner 324, and passed on to the corresponding switch(es)304. GaAs MESFET switches of the type well known in the art in0.35-micron technology are used, although other processes (such as 0.18micron or 0.1 micron) may be substituted. Device models correlated fromactual measured transistors at a GaAs semiconductor wafer foundry havealso been used herein as a simulation basis. Gallium arsenide (GaAs)MESFET or PHEMT switches are commonly used when a combination of speed,power, and efficiency requirements make them an attractive choice.However, the switches 304 of the illustrated embodiment may beimplemented in any one of many available technologies, the invention notbeing limited to GaAs switch technology. For example, complementaryoxide metal semiconductor (CMOS) switches meeting the speed, power, andefficiency requirements for a particular application may be useful, andalso desirable from a cost perspective.

Nor is the present invention limited in any way to FET device types. Forexample, a bipolar switch may be sufficient in some applications of theinvention in place of the FET. Accordingly, the switches 304 of FIG. 3are represented as simple ideal switch models.

While it is well known to one of ordinary skill in the art that GaAsMESFETs make efficient high speed power switches due to their inherenthigh electron mobility and other salient physical properties, it is alsorecognized that GHz-speed logic in CMOS is extremely power consumptiveif the outer limits of the technology are pushed, where current-modelogic is needed to meet the speed requirements. It will be appreciatedthat while GaAs MESFET logic has a much lower power-delay product thanCMOS, particularly at the most extreme end of the technology limits,GaAs technology is generally not considered a tenable choice for digitalprocessing logic (such as that of the exemplary processor 310 of FIG. 3)at the time of this invention disclosure. Additionally, takinghigh-speed clocks on and off chip at GHz speeds can be very powerconsumptive due to a high CV²F power loss factor. Heretofore, theseissues posed a difficult dilemma with no clear answer.

The benefits of the present invention would therefore be furtherleveraged through a switch solution that addressed these issues; i.e.,which further lowered power consumption and cost. One such power- andcost-efficient solution comprises integrating the encoder 302 andcharging switches 304 on one monolithic GaAs chip. One of ordinary skillin the art will appreciate that GaAs and other III-V compoundsemiconductors are well suited for RF power amplifiers and switchesoperating at RF speeds. Other semiconductor technologies may besimilarly suited, such as silicon germanium (SiGe) and indium phosphide(InP), and others may emerge over time that also provide RF logic andpower integration benefits. The lower-speed logic circuits of FIG. 3 or4, however, can readily be implemented in CMOS, while aparallel-to-serial interface and a serial-to-parallel interfacecombination can be utilized between the lower speed CMOS logic and thehigher speed GaAs encoder logic 302 and switches 304.

Hence, the present invention contemplates any variety of differentconfigurations, including notably use of “hybrid” GaAs and CMOSconfigurations (i.e., certain portions of the apparatus of FIG. 3 onGaAs and other portions of the apparatus in CMOS). For example, in oneembodiment, RF components including the noise shaping encoders 302,Sample-and-hold interpolators 318, digital quadrature modulator 316, andcharging switches 304 are disposed on one or more GaAs devices, whilethe data processor 310 and first interpolators 312 are disposed in aseparate CMOS device coupled to the GaAs device(s) viaparallel-to-serial and serial-to-parallel interfaces. Other arrangementusing GaAs and CMOS for various of the components of the apparatus 300may be used consistent with the invention.

The switches 304 may be configured in any number of standardconfigurations, like the push-pull configuration shown in FIG. 4. InFIG. 3, the switches 304 are shown coupled to a resonator 306 through abalun transformer 308, which is effectively adifferential-to-single-ended converter. The transformer 308 and/orresonator 306 may be implemented in many different ways, including butnot limited to magnetically coupled transformers, coupled microstrip orstripline transformers, coaxial ceramic resonators, or helicalresonators. The transformer 308 may also incorporate some or all of thedesired characteristics of resonator 306.

The center resonant frequency of the transformer 308 and resonator 306are both set at F_(c). The resonator is assumed to have a relativelyhigh unloaded Q-factor, so that its energy storage capability is highand very little power is wasted in the process. For example, if F_(c)equals 1.88 GHz, and if the desired bandwidth of the transmitter is 100MHz, then the loaded Q is 1880/100=18. If the efficiency loss of theresonator is 5%, then unloaded Q is 18/0.05=360. The resulting spectralinputs and outputs of the resonator are depicted in FIGS. 7( c), 8(c),and 9(c). The resonator effectively removes the out-of-band quantizationnoise from the noise-shaping encoders 302 to an acceptable level.

In another exemplary embodiment, the transformer 308 and resonator 306are combined with a helical resonator. Helical resonators are well knownin the art, and information thereon can be found for example inReference Data for Radio Engineers, Fifth Edition, copyright Howard W.Sams & Co. (ITT), pages 22-28 through 22-30. The coupling between theswitches 304 and the helical resonator 306 may be either probe, loop, oraperture coupling. On especially useful method of coupling is loopcoupling because both the phase and the anti-phase polarities may beeasily obtained, although it will be recognized that other methods maybe used with success in the invention.

FIG. 4 depicts in relevant part an equivalent circuit diagram of theswitch/transformer/resonator interfaces. The switches can be configuredas differentially charging the resonator's equivalent capacitor C_(T).The resonator's equivalent capacitor C_(T) is commutated (turned over inpolarity) by the resonator's equivalent inductor L_(T) during the timeperiods of the zero states. The coupling is depicted as a transformerwith turns ratio N. For the input coupling, the transformer is depictedas N_(i) with two opposite-phased primary windings and one secondarywinding, and for the output coupling, the transformer is depicted asN_(o) with one primary winding and one secondary winding. The effectiveturns ratios N_(k) transform the impedances as the square of the turnsratio such that the switches can supply the required amount of charge tothe C_(T) in the time that the switches are closed.

Following the resonator 306, any combination of elements, including butnot limited to, a lowpass filter, a transmit/receive (T/R) switch, or aduplexer, may be optionally employed before, or as part of, the antennaconnection 309. For example, FIG. 11 illustrates the incorporation ofthe transformer 308 and resonator 306 into a duplexer 1102 in anefficient manner that potentially saves cost and improves efficiency byreducing the number of separate elements needed to provide thefunctionality required in a given application.

Regarding the resonator and transformer bandwidths, by way of example,IS-95, IS-95a, IS-98, and IS-2000 include the North American PCStransmit bands covering 60 MHz, from 1.85-1.91 GHz. If the transformer308 and resonator 306 have a fixed tuning, it may be desirable to maketheir frequency response wider than 60 MHz for several reasons: (1) tokeep the insertion loss at the band edges at a minimum; and (2) to keepphase shifts and reflection coefficients from varying too much over thefrequency band of interest. In applications where a more narrowbandresonator is allowed, or where the resonator can be automatically tunedon the fly, these problems may be mitigated to some degree, and agreater amount of out-of-band rejection of quantization noise from thenoise-shaping encoders may result, providing a cleaner and more coherentoutput, and more efficiency from the switch delivering energy into thedesired band and less of the energy wasted in non-coherent excitation.However, a narrower resonator bandwidth requires a greater unloaded Q tokeep the insertion losses from being too great, which could partiallydefeat the benefits of added efficiency.

The transformer 308 and/or resonator 306 are charged positively (at thepeaks) and negatively (at the valleys) of the carrier frequency F_(c)every 180 degrees by the data coming out of the switches 304. Atime-domain plot (FIG. 12) illustrates this concept. This action ensuresthat energy is not wasted by charging the resonator at the wrong momentin time, but this most efficient action only occurs during periods whenthe amplitude and phase is very slowing changing, or when the encoderhas a high pulse density. A more rapid change of modulation or a lowpulse density results in more frequent anti-phasing or discharging ofthe resonator.

In some applications, amplitude and power level control can be performedin a purely digital fashion by simply changing the digital gain anywherein the digital data path. It may be accomplished at the lowest samplerate in the digital processor 310 prior to the interpolators 312.Alternatively, it may be accomplished during or after the interpolationfilter 312. For CDMA IS-95 or CDMA 2000, the output power level of thehandset must be able to vary over at least an 80 dB range.

There are at least two significant advantages to adding analog powercontrol as an extra degree of freedom to the digital power level scheme.Depending on the characteristics of the encoders 302, there may not beenough dynamic range to vary the output power purely in an all-digitalmanner and still meet the out-of-band energy suppression required by thesystem standard, especially in CDMA. Secondly, the efficiency of thesystem shown in FIG. 3 is greatly improved at lower power levels byallowing the DC bias to vary or step down at the lower power levels, andstill provide enough bias to keep the switch in a useful range ofoperation. The design of DC-DC converters is well known and commonlypracticed in the art, and therefore not further described herein.

In addition to DC bias control, a digitally controlled on-the-flydevice-size scaling of the switches may be used to provide additionalpower control. One may think of this as a semi-digital mechanism thatincorporates analog and digital aspects working together. Assuming theswitches are operating as current-limited devices rather thanvoltage-limited devices, they do not need to be as large as at the highpower levels, since the current of the switch is proportional to thedevice area. At lower power levels, less current is needed, hence, lessdevice area.

Therefore, with the DARP converter as disclosed herein, power gaincontrol may be purely digital or a combination of digital and analog,depending on the tradeoffs needed in the application.

It may be desirable to provide an on-the-fly dynamic impedancetermination during periods when both switches 304 a, 304 b are open,especially during times when long string of zeros is coming out of thedigital quadrature modulator 316. In this case, a separate terminationswitch 1002 (FIG. 10) can effectively switch on to activate atermination impedance network 1004, consisting of an imaginary part,which is a DC blocking capacitor, and a real part, some sort of resistorwhose value is approximately equal to the driving point impedancelooking into the transformer 308. The advantage of this terminationnetwork is that it keeps undesirable reflections from the load impedance309 from interfering with the ideally expected linear time-invariantbehavior of the resonator as seen by the charging switches 304.

While this invention disclosure has dealt with a rectangular orCartesian I-Q coordinate system representation, it will be obvious toone of ordinary skill in the art of digital communications systems thatthese same concepts can be implemented in polar form instead ofrectangular form, such that the I and Q vectors are converted into thepolar form of magnitude and phase at some point in the signal processingpath of the apparatus of FIG. 3 or 4. Also, it is known in the prior artthat noise-shaped encoding can be performed on the magnitude vector, andon the phase vector, rather than on the I and Q vectors only. It will beappreciated that the switches 304 can be driven from the quantizednoise-shaped phase information, and ‘envelope restoration’ techniquesknown in the art of Class-E amplifier design can be applied to modulatethe DC power supply containing the magnitude or envelope. The signalbandwidth of the envelope is approximately equal to the symbol bandwidthset by the symbol-rate interpolation filter described herein, so theenvelope information may vary at a slower rate than the phaseinformation. Hence, based on a combination of this invention disclosureand on the known art relating to efficient Class-E amplifier design andenvelope restoration techniques, that a specific alternative embodimentmay be readily obtained, and that such an alternate embodiment may havesome advantages over the rectangular coordinate-based examples detailedin this disclosure. Because such polar techniques are known in the art,they are assumed a fundamental alternative embodiment of this inventionas described above without the need for further detail or disclosureherein.

FIGS. 13-20 herein provide additional MATLAB simulation outputs ofvarious aspects of the invention.

FIG. 13 illustrates the spectral output of the final stage ofinterpolation filtering at 1536 times the symbol rate.

FIG. 14 illustrates the spectral output of the encoder 302, at the inputto the switches 304. The third-order modulator previously described inthe invention disclosure was used as the basis for generating thisoutput.

FIG. 15 illustrates the frequency response of the exemplary resonator306.

FIG. 16 illustrates the spectral output of the resonator 306 over thefull measurable band.

FIG. 17 illustrates the spectral output of the resonator 306 over thePCS transmit band.

FIG. 18 illustrates the transmit leakage power into the PCS receive bandwith an 80 MHz offset.

FIG. 19 illustrates the time domain output of the exemplary resonator306.

FIG. 20 illustrates the phase response of an ideal case withoutdelta-sigma modulation, and with delta-sigma modulation.

FIG. 21 illustrates the difference in the time domain obtained fromCadence (SPICE) simulations of the switch/resonator circuit performed bythe Assignee hereof, using actual production-quality GaAs MESFET models(from a wafer foundry), as compared to an ideal switch simulation.Actual CDMA waveforms pre-processed from a MATLAB, simulation of thedigital processing blocks in the invention that precedes theswitch/resonator interface, and the input stimulus for the switches,were imported into Cadence for simulation and analysis.

It will be recognized that while certain aspects of the invention aredescribed in terms of a specific sequence of steps of a method orordering of components in an apparatus adapted to implement themethodology of the invention, these descriptions are only illustrativeof the broader invention, and may be modified as required by theparticular application. Certain steps/components may be renderedunnecessary or optional under certain circumstances. Additionally,certain steps/components or functionality may be added to the disclosedembodiments, or the order of performance of two or more steps orcomponents permuted. All such variations are considered to beencompassed within the invention disclosed and claimed herein.

While the above detailed description has shown, described, and pointedout novel features of the invention as applied to various embodiments,it will be understood that various omissions, substitutions, and changesin the form and details of the device or process illustrated may be madeby those skilled in the art without departing from the invention. Theforegoing description is of the best mode presently contemplated ofcarrying out the invention. This description is in no way meant to belimiting, but rather should be taken as illustrative of the generalprinciples of the invention. The scope of the invention should bedetermined with reference to the claims.

1. An apparatus comprising: a pulse input source configured to generatea plurality of pulses, the pulse input source comprising a chargingswitch configured to sample at least one of a voltage and a current of apower supply and deliver the power supply samples to the firstresonator; the first resonator operatively coupled to the pulse inputsource and configured to receive input signals generated based on one ormore of the pulses; a second resonator operatively coupled to andconfigured to receive signals from an output of said first resonator andgenerate a second output to a receiver; and a load impedance disposedbetween said first and second resonators configured to facilitate atleast one of: transmission and reception of radio frequency (RF)signals.
 2. The apparatus of claim 1 further comprising: a transformercoupled to an input of the first resonator and adapted to generate theinput signals.
 3. The apparatus of claim 1, wherein the resonatorarchitecture comprises at least part of a direct-conversionarchitecture, and at least one of the first and second resonators iscapable of operating at or near a carrier frequency.
 4. The apparatus ofclaim 1, wherein the forms first resonator, the second resonator, andthe load impedance form a duplexer and wherein the first and secondresonators are capable of operating at or near a carrier frequency. 5.The apparatus of claim 1 further comprising: a transformer operativelycoupled to an input of the first resonator and an output of the pulseinput source, the transformer configured to generate the input signalsbased on one or more of the pulses.
 6. The apparatus of claim 1, whereinthe pulse input source comprises: a noise-shaping encoder configured toreceive and encode data, wherein the charging switch is configured toreceive the encoded data from the noise-shaping encoder.
 7. Theapparatus of claim 6 further comprising: a transformer operativelycoupled to an input of the first resonator and an output of the pulseinput source, the transformer configured to generate the input signalsbased on operation of the pulse input source.
 8. The apparatus of claim7, wherein the noise-shaping encoder is further configured to distributequantization noise substantially outside at least one frequency bandassociated with the receiver.
 9. A resonant power converter comprising:a pulse input source configured to generate a plurality of pulses, thepulse input source comprising a charging switch configured to sample atleast one of a voltage and a current of a power supply and deliver thepower supply samples to the first resonator; the first resonatoroperatively coupled to the pulse input source and having a resonantfrequency, the first resonator configured to receive input signals; asecond resonator operatively coupled to and configured to receivesignals from an output of the first resonator and generate a secondoutput to a receiver; and a load impedance disposed between the firstand second resonators configured to facilitate at least one of:transmission and reception of radio frequency (RF) signals.
 10. Theresonant power converter of claim 9, wherein the pulse input sourcecomprises: a noise-shaping encoder configured to receive and encodedata, wherein the charging switch is configured to receive the encodeddata from the noise-shaping encoder.
 11. The resonant power converter ofclaim 10, wherein: the noise-shaping encoder is configured to suppressquantization noise at least at one frequency.
 12. The resonant powerconverter of claim 10 further comprising: a transformer operativelycoupled to an input of the first resonator and an output of the pulseinput source, the transformer configured to generate the input signalsbased on operation of the pulse input source.
 13. The resonant powerconverter of claim 12, wherein the noise-shaping encoder is furtherconfigured to distribute quantization noise substantially outside atleast one frequency band associated with the receiver.
 14. The resonantpower converter of claim 9, wherein the noise-shaping encoder comprises:a plurality of low-pass encoders, the plurality of low-pass encodersconfigured to provide a band-pass functionality.
 15. The resonant powerconverter of claim 9, wherein the first resonator comprises a high-Qpassive resonator configured to substantially multiply the loadimpedance at the output of the first resonator as seen by an input tothe first resonator.
 16. The resonant power converter of claim 9,wherein the first resonator is further configured to selectivelyreinforce at least portions of a plurality of the generated pulses so asto optimize an efficiency of the resonant power converter.
 17. Theresonator architecture of claim 9, wherein the first and secondresonators and the load impedance form a duplexer and wherein the firstand second resonators are capable of operating at or near a carrierfrequency.
 18. A method of processing data in a radio frequency (RF)communications system, the method comprising: generating a plurality ofpulses using a pulse input source; sampling, using a charging switch, atleast one of a voltage and a current of a power supply; delivering thepower supply samples to a first resonator; receiving input signals atthe first resonator operatively coupled to the pulse input source;receiving signals from an output of said first resonator and generatinga second output to a receiver, using a second resonator operativelycoupled to the first resonator; and at least one of transmitting andreceiving RF signals using an antenna disposed between the first andsecond resonators.
 19. The method of claim 18, wherein generating theplurality of pulses further comprises: receiving and encoding data usinga noise-shaping encoder.
 20. The method of claim 19 further comprising:generating the input signals based on operation of the pulse inputsource using a transformer operatively coupled to an input of the firstresonator and an output of the pulse input source.